A High-Throughput VLSI Architecture for Real-Time Optical OFDM Systems With an Efficient Phase Equalizer

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    A High-Throughput VLSI Architecture forReal-Time Optical OFDM SystemsWith an Efficient Phase Equalizer

    Architecture dITGE haut-dbit pour lessystmes optique temps rel OFDM avec

    un galiseur de phase efficaceReza Ghanaatian, Mahdi Shabany, and Morteza H. Shoreh

    Abstract In this paper, a novel high-throughput very large scale integrated circuit architecture for areal-time implementation of intensity modulation direct detection optical orthogonal frequency divisionmultiplexing system is proposed, achieving the highest throughput reported to date. The proposedarchitecture utilizes a fast, pipelined, and parallel inverse fast Fourier transform/fast Fourier transformin the transmitter/receiver, which is customized to satisfy the throughput requirements of the advancedoptical systems. In addition, an efficient high-accuracy equalization method is developed, improving thesystem performance compared with the conventional linear equalizers. To evaluate the system performance,the OptiSystem software is used to model the optical channel and a Virtex-6 ML-605 evaluation boardis used as the implementation platform. Moreover, the synthesis results in a 180-nm CMOS technologyprove that the proposed architecture achieves a sustained throughput of 22.5 Gb/s with a 4.89-mm2 corearea.

    Rsum Dans ce papier, une architecture de circuit intgr trs grande chelle (ITGE) et haut dbit estpropose pour une nouvelle mise en uvre dun systme temps rel de modulation dintensit optiqueorthogonale pour la dtection directe de la frquence des systmes de multiplexage, permettant ainsidatteindre de plus haut dbit jusqu nos jour. Larchitecture propose utilise une transforme de Fourierrapide (Fast Fourier Transform, FFT) inverse, en pipeline, et parallle dans lmetteur/rcepteur. Cettedernire a t adapte afin de satisfaire les contraintes de dbit des systmes optiques de pointe. galement,un procd dgalisation de haute prcision efficace est dvelopp, ce qui amliore les performances dusystme par rapport aux galiseurs linaires classiques. Pour valuer les performances du systme, lelogiciel OptiSystem est utilis pour modliser le canal optique et une carte dvaluation Virtex-6 ML-605est utilise comme plate-forme de dveloppement. De plus, les rsultats de la synthse avec une technologieCMOS de 180 nm prouvent que larchitecture propose permet dobtenir un dbit soutenu de 22.5 Gb/savec une puce microlectronique de 4.89 mm2.

    Index Terms Efficient phase equalizer, intensity modulation direct detection (IMDD), optical orthogonalfrequency-division multiplexing (OOFDM), throughput, very large scale integrated circuit (VLSI)architecture.


    ORTHOGONAL frequency-division multiplexing(OFDM) is widely used in both wired and wirelesssystems due to its various advantages, such as the spectralefficiency, great performance in multipath fading channels,and the simple hardware implementation [1]. Recently, ithas become the technology of choice for systems employingoptical communications [2]. The tremendous increase in

    Manuscript received August 26, 2013; revised January 6, 2014 and March 2,2014; accepted April 3, 2014. Date of current version August 15, 2014.

    The authors are with the Department of Electrical Engineering,Sharif University of Technology, Tehran 11369, Iran (e-mail:reza.ghanaatian@gmail.com; mahdi@sharif.edu; m.h.shoreh@gmail.com).

    Associate Editor managing this papers review: Reza Heidari.Color versions of one or more of the figures in this paper are available

    online at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/CJECE.2014.2317756

    demand for the network capacity due to the advent ofnew Internet applications, and the development in digitalsignal processing (DSP) technology, which enables theimplementation of sophisticated OFDM signal processingalgorithms, has created a great motive to use OFDM inoptical communication systems [3].

    Among optical OFDM (OOFDM) systems, the coherentoptical OFDM provides the ultimate performance on thereceiver sensitivity, spectral efficiency, and robustness againstdispersion [4]. On the other hand, the direct detection opti-cal OFDM (DD-OOFDM) systems come with a lower cost,appealing to various applications [4]. Intensity modulationdirect detection OOFDM (IMDD-OOFDM) is one type ofDD-OOFDM system, which is a promising solution towarddeveloping high-bandwidth access networks. This is mainlybecause of the fact that IMDD-OOFDM offers a reasonable

    0840-8688 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.


    reduction in the overall network complexity. Meanwhile, theexperimental demonstration of a real-time IMDD-OOFDMtransceiver is vital to enable the practical realization of theOFDM systems in next-generation optical networks. How-ever, the implementation of an OOFDM system has variouschallenges including devising a high-throughput hardwarearchitecture, which could handle the parallel, complex, andcomputationally intense OFDM signal processing algorithms.The real-time implementation of OOFDM systems has beenpreviously studied in [5][7]. However, an efficient architec-ture for the baseband part of the system at extremely high datarates is still a major challenge.

    In this paper, a real-time, high-throughput very largescale integrated circuit (VLSI) implementation of anIMDD-OOFDM transceiver is proposed, which provides greatperformance for the system via taking advantage of a highspeed, an efficient FFT, and a low-complexity symbol syn-chronizer in the transceiver. In addition to the above con-tribution, an efficient yet simple phase equalizer is devel-oped based on the nature of the optical channels, provid-ing better channel estimation. This effectively enhances thesystem performance compared with the conventional equal-ization approaches. The rest of this paper is organized asfollows. In Section II, the OOFDM system architecture isreviewed and the proposed hardware architecture for the real-time implementation is discussed. The transmission perfor-mance and the implementation results are demonstrated inSections III and IV, respectively. Finally, Section V concludesthis paper.


    A. System Model

    Fig. 1 shows the block diagram of a real-time DD-OOFDMtransceiver. On the transmitter side, the parallel input data aremapped onto the constellation points using an M-quadratic-amplitude modulation (QAM), and then fed to a 32-pointparallel IFFT core. To generate real valued samples for theintensity modulator, only half of the subcarriers are used fordata (one zero component at the dc subcarrier and othersfor data), while others are left for their complex conjugate.A cyclic prefix (CP) of eight samples is used resulting ina 40 samples OFDM symbol. The real OFDM electricalsignal is then fed into a MachZehnder modulator (MZM)to modulate the laser intensity. Following the modulator, anerbium-doped fiber amplifier (EDFA) is utilized to adjustthe optical launch power to measure the bit-error-rate (BER)performance. Finally, the optical signal is transmitted throughthe optical link (see Section III for more details).

    On the receiver side, an EDFA is used to amplify thereceived signal power followed by a simple photodetector toperform the direct detection. The samples of the convertedphotocurrent are then fed to the OFDM receiver, where thesymbol synchronizer, the FFT, the channel equalizer, and thedemodulator recover data in each subcarrier (Fig. 1).

    With respect to the architecture in Fig. 1, in this paper, theOFDM transmitter and receiver are implemented on a XilinxVirtex-6 Field-programmable gate array (FPGA) (see Section

    Fig. 1. Block diagram of the proposed real-time DD-OOFDM system.

    IV for details). The OptiSystem is used for the simulationof the optical channel and other optical devices used in thesystem. The OFDM symbols are saved and imported to thesoftware and the output of the channel is exported, and fed tothe FPGA.

    B. Optical OFDM System Description

    1) Symbol Synchronization: The symbol synchronizationis one of the essential blocks at the receiver in an OFDMsystem, which plays an important role in the overall systemperformance. For the purpose of the symbol synchronization,one task is to find the symbol start. The conventional methodto estimate the OFDM symbol start is using the correlationconcept. Considering an OFDM symbol y(n), the correlationof this signal with its shifted version can be defined as

    (n) =+Ng+1

    i=y(n + i) y(n + N + i) (1)

    where N represents the number of FFT points, Ng is thelength of the CP region, and shows the initial random offsetof the receiving OFDM samples. The result of the abovecorrelation is maximized when the correlation is calculatedin the CP region. This means by detecting the peak absolutevalue of , the beginning of the symbol can be found.

    It can be shown that the performance of the above methoddeteriorates in noisy environments as well as in cases wherethe FFT size is small, such as OOFDM systems. Therefore,in this paper, it is proposed to use a folding technique wherethe value of is added to its shifted version with the lengthof OFDM symbol as follows:

    k[n] = 1k


    i=0 [n i Ns] (2)

    for example, the five order of folding can be derived as

    5[n] = 15 { [n] + [n Ns] + [n 2Ns]+ [n 3Ns] + [n 4Ns]} (3)

    where Ns represents the length of OFDM symbol that is equalto N +Ng . In other words, the folding implies that the numberof samples in (1) increases, which leads to improving the SNRvalue. This is mainly because of the fact that, more than onesymbol is observed during the correlation, which results in animprovement in the algorithm accuracy. Fig. 2 shows and 5values calculated for three consecutive OFDM symbols. As itcan be seen, the peek positions of are not easily recognizable


    Fig. 2. Calculated values for three consecutive OFDM symbols. (a) value. (b) 5 value.

    Fig. 3. Received constellations for some of the subcarriers 1, 5, 9, 13, and 15 (left to right). I = inphase and Q = quadrature.

    while with the help of the folding technique, the peaks occurat correct positions with a sharp pattern.

    2) Channel Equalizer: The channel equalizer is used to mit-igate the fiber nonlinearity and phase modulation effects [8].In this paper, two methods of equalization are considered.

    a) Conventional linear equalizer: In conventional linearequalizers, a channel response for each OFDM subcarrier iscalculated based on the reference signals as follows [9]:

    Hk = XkXk,ref


    where Hk is the estimated channel response in the kth subcar-rier and Xk and Xk,ref are the received and the reference signalin the kth subcarrier, respectively. By multiplying the inverseof each subchannel response by the corresponding receivedsignal, the equalized subcarrier Yk is manipulated as follows:

    Yk = (Hk)1 Xk . (5)In other words, in this method, the calculation of the

    channel response for each OFDM subcarrier requires thecomputation of 2M parameters, where M is the number ofactive subcarriers.

    b) Proposed equalizer: In this paper, an efficient three-stage phase and amplitude equalizer/compensator is proposedwhose BER performance outperforms that of the conventionallinear equalizer. In this method, the equalizer function (inversechannel response) is represented by

    H1( f ) = H0e j H1( f ) (6)where H0 is the amplitude of the equalizer function. For thephase-shift keying scenario, since the data are modulated only

    on the phase of the signal, without loss of generality, it isassumed that H0 = 1. In addition, based on simulation results,as shown in Fig. 3, each subcarrier has a constant phase shiftrelative to its adjacent subcarriers. Thus, the inverse channelresponse is modeled by a linear subcarrier-dependent phaseshift as

    H1( f ) = k + (7)where and are constants, which will be discussed in thefollowing, and k is the subcarrier index such that f (k) =fc + k fsc, in which fc is the carrier frequency and fsc isthe subcarrier frequency spacing.

    In higher order modulations, such as 16-QAM and64-QAM schemes, the magnitude compensation is also nec-essary. Hence, the amplitude distortion of the received signalcan be compensated by H0 = e, in which is a constant andits calculation will be discussed in the following. As a result,the total equalization procedure is formulated as

    Yk = e+ j (k+) Xk (8)where Yk is the equalized signal and Xk is the receivedsignal. In summary, in the proposed equalization method inthis paper, the equalization process is simplified to calculatethree constants for all subcarriers. In other words, in theconventional linear equalizer, a subchannel response mustbe calculated for each subcarrier; however, in the proposedmethod, the equalization of all subcarriers is performed usingthree parameters of , , and . The calculation method ofthese three parameters are performed in three steps, explainedin the following.

    Step 1 ( calculation): To compute , a training sequenceis employed, and the BER is calculated for a range of


    varying . Then, the optimum value of , which minimizesthe BER value, is chosen. To achieve further performanceimprovement, this procedure is repeated for a shorter intervalaround the estimated . This approach continues until thecomputed satisfies the desired precision.

    Step 2 ( calculation): On the other hand, is chosen sothat the whole constellation is rotated to the position, thatthe average phase of the estimated constellation points Yn isequal to the phase of the transmitted constellation points Yn .Therefore, by sending Nt training symbols, will be definedas the geometric mean of the phase error as follows:

    = 1Nt


    n=1[arg(Yn) arg(Yn)] . (9)

    Fig. 4 shows the procedure for computation of and .Step 3 ( calculation): The parameter is chosen so that the

    whole constellation is placed at the position that the magnitudeof the estimated constellation points is equal to that of thereceived constellation points. Thus, by sending Nt trainingsymbols, will be defined as the geometric mean of theabsolute ratio between the amplitude of the estimated signalYn and the received signal Xn as follows:

    = ln







    . (10)

    It is interesting to note that the coefficients and are linearly dependent on the frequency spacing among thesubcarriers as well as the total length of the fiber used betweenthe transmitter and the receiver. Therefore, once their valuesare known for one setup, they can be easily calculated foranother setup with different length and frequency spacing bysimple arithmetic manipulations. Furthermore, considering thelong-t...